Method of operation of receiver combiner for spatial diversity digital communications

ABSTRACT

Apparatus for and method of processing received signals for use with multiple antenna spatial diversity receivers used in a digital communications system. The receiver has received signal phase modulator apparatus for each diversity received signal. The phase modulator provides a two state phase perturbation of zero or pi radians, based on the carrier frequency, which is switched to modulate the phase of the received signal for a fifty percent duty cycle over a symbol period.

FIELD OF THE INVENTION

[0001] This invention relates to communications systems and inparticular to a spatial diversity receiving system for receiving adigitally modulated signal.

BACKGROUND OF THE INVENTION

[0002] There are several digital modulation techniques, including BPSK,QAM, (sometimes referred to as QASK), QPSK, which can be demodulated bya delay and multiply operation known as differential detection. Inpoint-to-point and point-to-multi-point communications systems,particularly mobile radio systems, it is known that the medium's channelproperties vary over time. These channel variations cause the receiversignal to fade in and out or be subject to intermittent outages.Examples of media or communications channels exhibiting fadingproperties include electromagnetic signals propagating through theEarth's atmosphere, or undersea acoustic signals.

[0003] Signal fading in communications channels is a natural phenomenonthat limits the range of separation between the transmitter andreceiver. Many very practical and useful channels exhibit fading. Thefading in these channels is usually caused by the receiver being linkedto the transmitter by more than one propagation path, the lengths ofwhich vary with time. Such channels usually exhibit filing propertiestermined Rayleigh fadfing or Rician fading. Fading can also be caused bytime varying absorption. For example, the absorption properties of anatmospheric radio channel depend on the moisture content of theatmosphere, which changes over time.

[0004] The reception over a channel subject to fading, can be improvedby incorporating diversity to reduce the fading and intermittent channeleffects that interfere with communications over the channel. There areseveral diversity options including multiple transmissions using thesame communication frequency with each transmission separated in timefrom the other, which may be termed time diversity; multiple carrierfrequencies requiring several different communication frequencies, whichmay be termed frequency diversity; and multiple antennas or receivingelements with each antenna or receiving element physically separate fromthe other, which may be termed spatial diversity; etc.

[0005] It is well known that the reception of a signal that propagatesthrough a fading or intermittent channel can be improved through the useof spatial diversity. Spatial diversity systems utilize multiple,physically separated, receiving elements in a way that mitigates thefading or intermittent outages experienced by each of the receivingelements. A strong persistent signal, with a much smaller degree offading, is obtained by properly processing the signals from each of thereceiving elements and then properly combining these signals. Thestrength and persistence of the combined signal depends on the number ofreceiving elements and the techniques used to process and combine thesignals. There is a variety of signal processing and combiningtechniques that can be used.

[0006] One approach to reducing signal fading is to use some form ofdiversity to receive multiple signals at the receiver and then tocombine these signals in a constructive way. A diversity system couldinvolve frequency diversity which is implemented by transmittingidentical information on two or more separate carriers that areseparated enough in frequency for the fading of each carrier to beuncorrelated with the others. It could involve sending the identicalinformation on the same carrier but at two or more different times withenough separation in time for the fading at each time to be uncorrelated(time diversity). It could also involve sending the information onlyonce, on one carrier and receiving it on two or more separate antennasphysically separated with enough distance between them for the fading ateach antenna in the receiver to be uncorrelated with the others (spatialdiversity).

[0007] In all diversity systems the multiple signals have to becombined, which usually involves processing or conditioning each of themultiple signals and then summing or selecting these processed signals.There are different methods for combining the signals, each offeringdifferent trade offs between performance and implementation complexity.

[0008] Each of the three types of diversity has its advantages anddisadvantages. However all three types require a diversity combiner.Spatial diversity has the strong advantage of requiring minimalbandwidth to transmit the information. It has the disadvantage in radiofrequency communication of requiring more high frequency circuitry thanthe others, in particular multiple antennas and associated radiofrequency (RF) electronics. Spatial diversity is particularly attractivewhen the carrier frequency is sufficiently high for the antennas to beimplemented as printed circuits. At these frequencies the cost ofimplementing spatial diversity is the cost of the electronics associatedwith each antenna.

[0009] The theory of spatial diversity and its ability to mitigatefading is well known. To reach the theoretical performance limitrequires knowledge of the amplitude and phase of the carrier, whichchanges with time and can only be estimated. The challenge in achievingnear optimal performance is in the implementation of an amplitude andphase estimator and the implementation of the phase and amplitudecorrector. There are many techniques for estimating and correctingamplitude and phase and for combining the corrected signals. The varioustechniques have different degrees of compromise between performance andease of implementation. Examples of prior art that fall into thiscategory include:

[0010] U.S. Pat. No. 4,386,435 to Ulmer providing a space diversityreceiver includes an IF band combiner amplifier to sum the IF signalswhere one signal has a phase corrector which adjusts the phase inresponse to received signal characteristics. A similar approach is alsoused in U.S. Pat. No. 4,326,294 and U.S. Pat. No. 4,710,975 both toOkamoto et al.

[0011] U.S. Pat. No. 5,530,925 to Garner combines signals from twophysically diverse antennas after down conversion to an intermediatefrequency. U.S. Pat. No. 4,498,885 to Namiki combines two spatiallydiverse signals relying on controlled phase shifting of one of thesignals to cancel the effects of an interference wave.

[0012] In U.S. Pat. No. 5,203,025 to Anvari et al. the relative phaseand amplitude of the IF stage of a spatially diverse receiver are usedto determine a signal combining strategy of either direct summation, orinversion then summation, for recovering the modulating signal. Arelated approach is employed by Karabinis in U.S. Pat. No. 4,373,210which selects from two spatially diverse signals based on relativesignal to interference ratios. The receiver in U.S. Pat. No. 4,079,318to Kinoshita combines signals from spatially diverse antennas relying onphase control at the intermediate frequency stage.

[0013] There are several techniques for combining the signals frommultiple antennas to mitigate the effects of fading. These techniquescan be logically divided into two categories: post-detection combiningand pre-detection combining. The post-detection combiners essentiallyrequire an entire receiver for each antenna but the decision and controlcircuits remain common. While these circuits can achieve near optimumperformance, they are expensive solutions. Pre-detection schemes canalso be very complicated and expensive. To achieve near optimumperformance requires circuits that estimate the phase of the carrierreceived at each antenna as well as voltage controlled phase shiftercircuits to perform in phase alignment or correction. The purpose ofthese circuits is to align the phase of the carriers received on eachantenna. The implementation cost of such systems is quite high.

[0014] A simpler method that yields suboptimum performance can bedescribed as a frequency stacked IF combiner. In this method thecarriers from each antenna are translated to separate IF frequencies,with the separation between IF carriers being multiples of the bit rate.The multiplicity of frequency stacked IF signals is summed withoutregard to phase and then detected. This pre-detection method has thedisadvantage of requiring distinct electronics to convert from RF to IFfor each antenna, but avoids the need for estimation of carrier phaseand amplitude as well as avoids estimating the signal quality. Here thesignals from each receiving element are translated to distinct IF bands,the centers of which must be separated in frequency by multiples of thebit rate. The I.F. signals are combined and then detected by adifferential detector. This method is described in the paper by T.Masamura, “Frequency Offset Receiver Diversity for Differential MSK”,IEEE Trans. Veh. Technol., Vol. VT-36, No. 2, May 1987, pp. 63-70. Thistechnique is referred to as a frequency stacked IF combiner.

SUMMARY OF THE INVENTION

[0015] The present invention encompasses a method and preferredembodiment that provides for combining spatial diversity signals.

[0016] In accordance with the present invention, diversity signals fromreceiving elements are combined prior to detection obviating the needfor carrier signal recovery or detection prior to modulation signaldetection.

[0017] In another aspect of the present invention, spatially diversereceived signals are directly summed or combined without the need toestimate the phase of the signals from each receiving element, thuseliminating the requirements of the prior art apparatus and methods toobtain phase information and perform phase adjustments prior tocombining.

[0018] It is yet another aspect of the present invention that spatiallydiverse received signals are summed or combined without the need todetermine or estimate received signal quality of any of the signalreceiving elements to effect signal combining thereby eliminating theneed to estimate or determine signal strength or bit error rate for eachor any of the received signals.

[0019] In yet another aspect of the present invention, a receiver can beimplemented using a single down converter to an intermediate frequency(IF) stage providing a significant advantage in receiver constructionand economy over receivers requiring multiple IF stage receivers.

[0020] The receiver of the present invention includes arrangementsincluding multiple diverse receiving elements and is not limited to tworeceiving elements.

[0021] An object of the present invention is to substantially reducesignal fading in a receiver through the economical implementation ofspatial diversity receiving elements.

DESCRIPTION OF THE INVENTION

[0022] The present invention includes a method and apparatus to carryout the method and has the advantage of simple implementation. Themethod comprises perturbing the phase of the carrier of a receivedsignal from a diversity receiving element such that the perturbation ofthe phase of the carrier is periodic with a period equal to, or amultiple of, the symbol rate. For a plurality of diversity receivingelements, each of the periodic phase perturbations is different.Improved performance in accordance with the invention is obtained whenthe phase perturbations of the diversity receiving elements are mutuallyorthogonal over a symbol period. The perturbed phase signals are summedand then detected using a suitable detector. The detector can be, forexample, a differential detector.

[0023] In a preferred embodiment, binary phase perturbations are appliedto each received signal by switching the signal between two circuitpaths, one of which has more delay due to extra length. The differencein delay between the two paths is one-half of the carrier cycle. Thecircuit path switches are controlled by a set of binary functions thatare mutually orthogonal over one symbol period (for example, over onedata bit interval for coding schemes providing a single bit per symbol)and are each periodic with a common period equal to the symbol period.The binary functions are generated from a square clock using standardlogic elements. While there are many sets of binary functions that willwork, a set of functions known as Walsh functions are advantageouslyemployed to achieve the advantages of the present invention based onbinary phase perturbations.

[0024] The present invention discloses a method to combine the signalsfrom each antenna prior to down conversion. In the preferred embodiment,the phase perturber and combiner are placed immediately after theantennas so that only one low noise amplifier is required. In analternate embodiment, the phase perturber and combiner are placed aftera set of low noise amplifiers. Placement of the phase perturber andcombiner after low noise amplification of each received signal increasesreceiver cost but decreases the noise figure of the receiver system.

[0025] The method of the invention involves perturbing or modulating thephase of the carriers from each antenna in a periodic manner withoutregard for the phase relationship among the carriers received at thedifferent antennas. The method will reduce signal fading as long as thephase of the carriers are perturbed differently and the period of theperturbations is substantially the same as the symbol rate or a multiplethereof. The maximum possible reduction in fading is obtained if thephase perturbations of the carriers are orthogonal providing themagnitude of the perturbation is sufficient.

Theory of Operation

[0026] It is believed that the invention works according to thefollowing theory. The invention is not to be limited by the theory, butrather resides in the apparatus and a series of steps comprising themethod.

[0027] In accordance with the theory, a signal received at any givenreceiving element i in a plurality of receiving elements as function oftime may be expressed as:

r _(i)(t)=A _(i) cos(ω₀ t+Φ _(i))   (1)

[0028] where:

[0029] A_(i) is the amplitude of the carrier received at receivingelement i

[0030] ω₀ is the angular frequency of the carrier

[0031] Φ_(i) is phase in radians of the carrier received at receivingelement i. Where the channel linking the receiving element to thetransmitter is slowly fading, the carrier amplitude and phase areessentially constant for the duration of two symbols. Thus the phaseΦ_(i) will be substantially constant over 2 successive symbol periodsfacilitating differential detection.

[0032] When receiver diversity is employed, there are signals from Nreceiving elements which may be summed to produce a combined signalwhich is given by:${s(t)} = {\sum\limits_{i = 1}^{N}{A_{j}\cos \quad ( {{\omega_{0}t} + \varphi_{j}} )}}$

[0033] The energy of a symbol in the combined signal is given by:E_(s) = ∫₀^(T)s²(t)t  where  T  is  the  symbol  period

[0034] Substituting for s(t) and expressing the square of a sum as adouble sum, the above integral may be used to express the combinedsymbol energy of all signals from N receiving elements as follows:$E_{s} = {\int_{0}^{T}{\sum\limits_{j = 1}^{N}{\sum\limits_{i = 1}^{N}{A_{i}A_{j}\cos \quad ( {{\omega_{0}t} + \varphi_{i}} )\cos \quad ( {{\omega_{0}t} + \varphi_{j}} ){t}}}}}$

[0035] Given the trigonometric identity:cos(a)cos(b)=½└cos(a+b)+cos(a−b)┘, the previous equation can beexpressed as:$E_{s} = {\int_{0}^{T}{\sum\limits_{j = 1}^{N}{\sum\limits_{j = 1}^{N}{{\frac{A_{i}A_{j}}{2}\quad\lbrack {{\cos \quad ( {{2\omega_{0}t} + \varphi_{i} + \varphi_{j}} )} + {\cos \quad ( {\varphi_{i} - \varphi_{j}} )}} \rbrack}{t}}}}}$

[0036] For communications signals, the carrier frequency is selected tobe much, much greater than the symbol rate. This may be expressedmathematically as: ω₀>>2π/T. Thus, over a symbol period, integration ofthe term involving cos(2ω₀t . . . ) will go to zero. Changing the orderof summation and integration of the foregoing formula and recognizingthat for i=j, cos(Φ_(i)−Φ_(j))=cos (0)=1, allows E_(s) to be expressedas: $\begin{matrix}{E_{s} = {{\sum\limits_{i = 1}^{N}{\int_{0}^{T}\frac{A_{i}^{2}{t}}{2}}} + {\sum\limits_{j = 1}^{N}{\sum\limits_{\underset{j \neq j}{i = 1}}^{N}{\int_{0}^{T}{\frac{A_{i}A_{j}}{2}\cos \quad ( {\varphi_{i} - \varphi_{j}} ){t}}}}}}} & (2)\end{matrix}$

[0037] It is noted that there are two summands provided in thisequation. The summand for the single sum is independent of carrier phaseand is always positive irrespective of the receiving element i receivingthe signal. The summand for the single sum increases with increasing N.Thus, the received signal strength will increase as N increases. Whileeach A_(i) will vary with time, the probability that all A will be smalldrasically goes down as N increases.

[0038] On the other hand, the summand for the double sum is a functionof the received signal carrier phases and could be negative for one ormore combinations of i and j. The value of each term in the double sumdepends on the phase difference between the respective carrier phases Φiand Φj. Accordingly, the carrier phases could, and at times would, besuch that the double sum could contain a sufficient number of negativeterms which provide a sum that completely counteracts the single sumthereby forcing Es to zero, or some value near zero. When Es approacheszero, by definition, signal fading or signal loss results. Therefore,constraining the terms of the double sum to provide a total whichapproaches zero, will operate to counteract the negative effect thedouble sum has on the single sum thereby reducing or eliminating fadingand signal loss.

[0039] Phase modulation apparatus can be provided to modify or adjustthe phase of the received signal. This can be done by introducing a timevarying delay to the received signal. The received signal represented byequation (1) can be rewritten to include the time varying phase functionas follows:

r _(i)(t)=A _(i) cos(ω₀ t+Φ _(i)+Ψ_(i)(t))

[0040] where:

[0041] Ψ_(i)(t) expresses the instantaneous phase adjustment for thesignal received at each receiving element i as a function of time wherea time varying phase adjustment is introduced.

[0042] When the received signal for each receiving element i isindividually phase adjusted, it can be shown that the combined symbolenergy of all signals from N receiving elements, as previously given inequation (2), can then be represented by: $\begin{matrix}{E_{s} = {{\sum\limits_{i = 1}^{N}{\int_{0}^{T}\frac{A_{i}^{2}{t}}{2}}} + {\sum\limits_{j = 1}^{N}{\sum\limits_{\underset{j \neq j}{i = 1}}^{N}{\int_{0}^{T}{\frac{A_{i}A_{j}}{2}\cos \quad ( {\varphi_{i} - \varphi_{j} + {\psi_{i}(t)} - {\psi_{j}(t)}} ){t}}}}}}} & (3)\end{matrix}$

[0043] By examination of equation (3), it is observed that byintroducing individual phase adjustment of the received signal from eachreceiving element i, the summand for the double sum becomes a functionof both the individual received signal carrier phases, φ_(i) and φ_(j),and the individual signal phase adjustment functions, Ψi(t) and Ψj(t).

[0044] The invention comprises utilizing a family of functions,preferably Walsh functions, that may be employed advantageously toprovide individual signal phase adjustment which operates to constrainthe double sum to approach zero for all permutations of the signalcarrier phases φ_(i)−φ_(j). Use of this family of functions operates tominimize the counteracting effect the double sum has on the single sumthereby reducing or eliminating fading and signal loss.

[0045] In accordance with the invention, a family of mutually orthogonalperiodic functions has the following properties:

[0046] 1. A first function has a single output state which is constant,either: high or low, one or zero etc. This can be expressed as follows:$f_{0} = \{ \begin{matrix}{1;} & {{{for}\quad 100\% \quad {of}\quad {the}\quad {time}};{or}} \\{0;} & {{for}\quad 100\% \quad {of}\quad {the}\quad {time}}\end{matrix} $

[0047] 2. For any other order function (f₁, f₂, f₃ . . . f_(N)), theoutput state will be high for one-half the time and low for one-half thetime. This may be expressed mathematically as follows:$f_{i} = \{ \begin{matrix}{1;} & {{{for}\quad 50\% \quad {of}\quad {the}\quad {time}};{and}} \\{0;} & {{for}\quad 50\% \quad {of}\quad {the}\quad {time}}\end{matrix} $

[0048] 3. Between any two higher order functions, over a given timeperiod, the output states of each individual function will be equalone-half the time and opposite one-half the time. This may be expressedmathematically as follows: ${f_{i}:f_{j}} = \{ \begin{matrix}{{0:0};} & {{{for}\quad 25\% \quad {of}\quad {the}\quad {time}};} \\{{0:1};} & {{{for}\quad 25\% \quad {of}\quad {the}\quad {time}};} \\{{1:0};} & {{{for}\quad 25\% \quad {of}\quad {the}\quad {time}};{and}} \\{{1:1};} & {{for}\quad 25\% \quad {of}\quad {the}\quad {time}}\end{matrix} $

[0049] The invention provides that such functions be used to producetime varying phase perturbations of the received signal for eachreceiving element i. The resulting time varying phase perturbation maybe expressed as: Ψ_(i)(t). The differences in the perturbated phasesbetween any two received signals can then be expressed as:Ψ_(i)(t)−Ψ_(j)(t). These functions are selected to be periodic over areceiver's symbol period. When such functions are so employed, theindividual phase perturbations of each received signal can be expressedas follows:

[0050] 1. Phase perturbation of a received signal i controlled byfunction f₀ may be expressed as a constant as follows:

Ψ_(i)(t)=Ψ_(i) for 100% of a symbol period, where Ψ_(i) could be 0  (P1)

[0051] 2. Phase perturbation of a received signal i controlled by anyfunction, other than f₀, may be expressed as: $\begin{matrix}{{\psi_{i}(t)} = \{ \begin{matrix}{0;} & {{{for}\quad 50\% \quad {of}\quad {the}\quad {time}};{and}} \\{\psi_{i};} & {{for}\quad 50\% \quad {of}\quad {the}\quad {time}}\end{matrix} } & ({P2})\end{matrix}$

[0052]  In accordance with the condition specified in P2, the phaseperturbation of any individual signal occurs for one-half of the time,or, stated another way, is subject to a fifty percent duty cycle. Forthe other one-half of the time, no phase adjustment of the signaloccurs.

[0053] 3. Phase perturbation phase differences between a received signali having a phase perturbation controlled by f₀ and any other receivedsignal j having a phase perturbation controlled by a function which isnot f₀, may be expressed as: $\begin{matrix}{{{\psi_{i}(t)} - {\psi_{j\quad}(t)}} = \{ \begin{matrix}{\psi_{i};} & {{for}\quad 50\% \quad {of}\quad a\quad {symbol}\quad {period}} \\{{\psi_{i} - \psi_{j}};} & {{for}\quad 50\% \quad {of}\quad a\quad {symbol}\quad {period}}\end{matrix} } & ({P3})\end{matrix}$

[0054] 4. Phase perturbation phase differences between any two signals iand j each having phase perturbations controlled by different functions,neither of which is f₀, may be expressed as: $\begin{matrix}{{{\psi_{i}(t)} - {\psi_{j}(t)}} = \{ \begin{matrix}O & {;{{for}\quad 25\% \quad {of}\quad a\quad {symbol}\quad {period}}} & (a) \\\psi_{i} & {;{{for}\quad 25\% \quad {of}\quad a\quad {symbol}\quad {period}}} & (b) \\{- \psi_{j}} & {;{{for}\quad 25\% \quad {of}\quad a\quad {symbol}\quad {period}}} & (c) \\{\psi_{i} - \psi_{j}} & {;{{for}\quad 25\% \quad {of}\quad a\quad {symbol}\quad {period}}} & (d)\end{matrix} } & ({P4})\end{matrix}$

[0055]  In accordance with the condition specified in P4, the phaseperturbation between any two phase perturbed signals has 4 possibleoutcomes each of which occurs equally, exactly 25% of the time in onesymbol period. Condition P2 still applies to each individual signal andrequires each to be phase perturbed for one half of the time. In otherwords, subject to a fifty percent duty cycle. Condition P4 requires thatthe phase perturbation between any two signals be mutually absent fortwenty-five percent of the time, mutually present for twenty-fivepercent of the time, and present in one but not the other fortwenty-five percent of the time and present in the other but not the onefor twenty-five percent of the time. For the purposes of the inventionherein described, any two binary functions are said to be mutuallyorthogonal when they meet the requirements of condition P4.

[0056] Without loss of generality, it can be assumed that functionΨ₁(t), which represents the phase adjustment of a first received signal(i.e. i=1), is a constant corresponding to function f₀. Accordingly,Ψ₁(t)=Ψ₁. This meets the requirements of the condition expressed inequation P1. Further, and again without loss of generality, it can beassumed that Ψ₂(t) through Ψ_(N)(t), which represent the phaseadjustment of the second through N received signals, is given by thefunctions f₁, f₂, . . . , f_(N−1). Accordingly, Ψ_(i)(t)=Ψ_(i) for 50%of a symbol time and Ψ_(i)(t)=0 for 50% of a symbol time. This meets therequirements of the condition expressed in equation P2.

[0057] Based on the simplifying assumptions of the previous paragraph,and incorporating the individual received signal phase perturbationsdefined by equations P3 and P4, the double sum summand expressed inequation (3) can be expressed as: $\begin{matrix}{E_{~\varphi} = {{2{\sum\limits_{j = 2}^{N}\quad {\int_{0}^{\frac{T}{2}}{{\frac{A_{1}A_{j}}{2}\quad\lbrack {{\cos \quad ( {\varphi_{1} - \varphi_{j} + \psi_{1}} )} + {\cos \quad ( {\varphi_{1} - \varphi_{j} + \psi_{1} - \psi_{j}} )}} \rbrack}{t}}}}} + {\sum\limits_{j = 2}^{N}\quad {\sum\limits_{\underset{i \neq j}{i = 2}}^{N}\quad {\frac{A_{i}A_{j}}{2}{\int_{0}^{\frac{T}{4}}{\lbrack {{\cos \quad ( {\varphi_{i} - \varphi_{j}} )} + {\cos ( {\varphi_{i} - \varphi_{j} + \psi_{i}} )} + {\cos ( {\varphi_{i} - \varphi_{j} - \psi_{j}} )} + {\cos \quad ( {\varphi_{i} - \varphi_{j} + \psi_{i} - \psi_{j}} )}}\quad \rbrack {t}}}}}}}} & (4)\end{matrix}$

[0058] where E_(φ) is used as a symbol for the phase dependent doublesum term, being the second summand or double sum of equation (3)

[0059] Given the trigonometric identity:

cos(a+π)=cos(a−π)=−cos(a),   (5)

[0060] it will be understood that the carrier phase dependent term,E_(φ) of equation (4), is forced to zero for any value of Ψ₁ if:

Ψ_(i)=π for i=2, . . . , N

[0061] For clarity, substitution of π for Ψ_(i) in equation (4) resultsin:$E_{~\varphi} = {{2{\sum\limits_{j = 2}^{N}\quad {\int_{0}^{\frac{T}{2}}{{\frac{A_{1}A_{j}}{2}\quad\lbrack {{\cos \quad ( {\varphi_{1} - \varphi_{j} + \psi_{1}} )} + {\cos \quad ( {\varphi_{1} - \varphi_{j} + \psi_{1} - \pi} )}} \rbrack}{t}}}}} + {\sum\limits_{j = 2}^{N}\quad {\sum\limits_{\underset{i \neq j}{i = 2}}^{N}\quad {\frac{A_{i}A_{j}}{2}{\int_{0}^{\frac{T}{4}}{\lbrack {{\cos \quad ( {\varphi_{i} - \varphi_{j}} )} + {\cos ( {\varphi_{i} - \varphi_{j} + \pi} )} + {\cos ( {\varphi_{i} - \varphi_{j} - \pi} )} + {\cos \quad ( {\varphi_{i} - \varphi_{j} + \pi - \pi} )}}\quad \rbrack {t}}}}}}}$

[0062] Performing the substitutions of the trigonometric identity givenin equation (5) results in: $\begin{matrix}{E_{~\varphi} = {{2{\sum\limits_{j = 2}^{N}\quad {\int_{0}^{\frac{T}{2}}{{\frac{A_{1}A_{j}}{2}\quad\lbrack {{\cos \quad ( {\varphi_{1} - \varphi_{j} + \psi_{1}} )} - {\cos \quad ( {\varphi_{1} - \varphi_{j} + \psi_{1}} )}} \rbrack}{t}}}}} + {\sum\limits_{j = 2}^{N}\quad {\sum\limits_{\underset{i \neq j}{i = 2}}^{N}\quad {\frac{A_{i}A_{j}}{2}\quad {\int_{0}^{\frac{T}{4}}{\lbrack {{\cos \quad ( {\varphi_{i} - \varphi_{j}} )} - {\cos ( {\varphi_{i} - \varphi_{j}} )} - {\cos ( {\varphi_{i} - \varphi_{j}} )} + {\cos \quad ( {\varphi_{i} - \varphi_{j}} )}}\quad \rbrack {t}}}}}}}} & (6)\end{matrix}$

[0063] By inspection of either equation (4) or (6), it is furtherunderstood that E_(φ) is forced to zero for any values of Ψ₁. Thus, withΨ_(i)=π for i=2, 3, . . . N, the cosine terms in the integrands sum tozero. This makes the integrals for all i, j equal to zero and therebyforces E_(φ) to zero. This is the desired objective.

[0064] A preferred embodiment of the present invention will now bedescribed with reference to the drawings in which like features of theinvention bear like reference numerals throughout the various figures.

BRIEF DESCRIPTION OF THE DRAWINGS

[0065]FIG. 1 is a functional block diagram of a first embodiment of aspatial diversity receiver in accordance with the invention.

[0066]FIG. 2 is a functional block diagram of the phase perturber andsignal combiner of FIG. 2.

[0067]FIG. 2a is a functional block diagram of an alternate embodimentthe phase perturber and signal combiner of FIG. 2.

[0068]FIG. 3 shows timing diagrams for clocking phase perturbations forup to 8 diversity receiver signals.

DESCRIPTION OF THE PREFERRED EMBODIMENT

[0069] Reference is made to FIG. 1 which shows a functional blockdiagram of a spatial diversity receiver incorporating a phase perturberand signal combiner of the present invention. A plurality of receivingelements 10 is provided, individually numbered for convenience as: 0, 1,2, 3, . . . N. It will be understood that a two antenna spatialdiversity receiver will employ two antennas, a three antenna spatialdiversity receiver three antennas and so on. Each of the individualantennas or receiving elements is physically displaced from one anotherto provide a received signal on each receiving element that does notcorrelate to or has minimal correlation with the signal received on anyother receiving element. That is to say, each received signal has asmall probability of simultaneously fading with another signal. Thegeneral, case is shown in FIGS. 2 and 3 where N receiving elements aredepicted. The output from each receiving element 10 is supplied to aphase perturber and signal combiner 12 which processes the signal inaccordance with the invention as more particularly described withreference to FIG. 2. Each signal is received on a receiving element andmay be amplified prior to being supplied to the phase perturber andsignal combiner using a suitable low noise amplifier (LNA) 11 to reducethe adverse effects of switching noise introduced by combiner 12. Theuse of individual LNA 11 for each receiving element is optional and willdepend on the constraints (e.g. transmitter power, battery energyconsumption etc.) inherent in the nature of the application in which areceiver, in accordance with the invention, is to be used. Whether thereceived signal is amplified upstream or downstream from the combiner 12does not affect the principles of operation of the present invention.However, amplifier placement upstream of the phase perturber and signalcombiner 12, as depicted in FIG. 1, will decrease the effects ofswitching noise and improve the noise figure of merit for the receiver.

[0070] The conditioned signal output from the phase perturber and signalcombiner 12 in accordance with the invention is provided on line 14. Theoutput conditioned signal is subsequently processed by the receiver in aconventional manner to recover the data encoded in the transmittedsignal. It will be understood that the signal processing required torecover the transmitted data will depend on the nature of the encodingof that data which was carried out at the transmitter. An exemplaryreceiver detector system is shown in FIG. 1 for processing theconditioned signal appearing at line 14.

[0071] In radio frequency (RF) systems, RF processing generally includesa frequency translation stage 16 to translate the conditioned signal toan intermediate frequency for data recovery. For digital radiocommunications, there are generally two types of detectors, coherentdetectors and non-coherent detectors. The differential detector is anon-coherent type of detector. The following example provides a summarydescription for a receiver based on differential detection which isemployed to detect symbols encoded without the need to recover a carriersignal as part of the symbol detection process. Following frequencytranslation at the frequency translation stage 16, the frequencytranslated signal is provided to a differential detector 18 to recoverthe data signal from the frequency translation stage. The output of thedifferential detector 18 is supplied to a matched filter 20 whichoperates co-operatively with the differential detector to produce anoutput level representative of the recovered symbol. It will beunderstood by those skilled in the art that the form of the matchedfilter and differential detector pair will naturally depend on thenature of the coding scheme employed for transmitting the informationover the radio channel. Any suitable digital differential coding schememay be readily used in conjunction with the signal processing andapparatus of the present invention and may include, by way ofillustration and exemplification such schemes as BPSK, QAM (or QASK asit is often called) and QPSK among others. Similarly, the signalcombining and phase perturbation teachings of the present invention mayalso be employed in coherent detector transmission schemes. However,persons skilled in the art will appreciate that the carrier recoveryapparatus of the receiver will need to contend with the phaseperturbations introduced by the signal combiner and perturber 12 thusrequiring multiple carrier recovery circuits. Notwithstanding this, thepresent invention may be economically and advantageously employed forcoherent detectors relying on embedded pilot signals, such as areemployed in spread spectrum transmission systems for example.

[0072] The output from the matched filter 20 is supplied to a data clockrecovery circuit 22 and a sample and hold circuit 26. The data clockrecovery circuit 22 is used to recover a data clock signal from thereceived signal to synchronize with transmitter symbol encoding tofacilitate symbol recovery by the receiver. The recovered data clocksignal is supplied by clock line 30 for use in bit recovery in thereceiver and also to provide the symbol period to drive phase perturberand signal combiner 12. In addition to providing a signal for clockrecovery, the matched filter 20 output is provided to a sample and holdcircuit 26 which samples the output signal at the appropriate time,based on the coding scheme and detector used, to produce a receiverestimate of the transmitted symbol. The appropriate time for filtersampling is determined by the data clock recovery clocking and, as iswell known to those skilled in the art, is preferably done near the endof the symbol time.

[0073] The sampled output is then provided to a quantizer 24 which is adecision box form of circuit, sometimes referred to as a discriminator.Quantizer 24 performs a quantization of the sampled signal to produce arecovered data output 28 that conforms to the bit encoding of the symbolin accordance with the transmission scheme employed. For example, abinary symbol providing for one bit of data per symbol might employ athreshold detector to discriminate between a positive or negativesampled signal to produce a normalized level output on recovered dataline 28 representative of a 1 or a 0. Multi-bit symbol encoding schemeswould similarly provide outputs representative of the 2-bit, 3-bit,n-bit bit tuples encoded in the symbol.

[0074]FIG. 2 shows a functional block diagram of the elements of thephase perturber and signal combiner 12 of FIG. 1. For simplicity, thereceived signal amplifiers (LNAs 11) have not been shown. However, itwill be understood that signal pre-amplification can occur for eachindividual signal supplied by receiving element 0, 1, 2, 3, . . . N.That is to say, pre-amplification of each individual signal, if done, isperformed before the signal is supplied to the phase perturber andsignal combiner 12.

[0075] Data clock line 30 is supplied to an orthogonal functiongenerator 36, for example a Walsh function generator, which includes aclock multiplier to create a symbol synchronous clocking signal that isa multiple of the data clock. The function generator 36 internal clockmultiplier receives a recovered data clock signal on line 30 which isused to create multiples of the recovered clock. Function generator 36produces output clocking signals on control lines 38 a, . . . 38 n tocontrol corresponding signal path selection switches 40 a, 40 b, . . .40 n. The number of output clocking signals produced by functiongenerator 36 will depend on the number of diversity receiving elementsdisposed in the receiver embodiment. The function generator outputclocking signal for any individual path switch 40 a, 40 b, . . . or 40 nis explained with reference to the clock timing diagrams of FIG. 3 anddescription accompanying that figure. The multiplied clock signal isused to produce mutually orthogonal function outputs (e.g. Walshfunctions) on each control line 38 _(N) that is used to control signalpath selection switches 40 _(N). For generality, N receiving elementshave been depicted in the phase perturber and signal combiner 12.However, it will be understood that if there are only 2 receivingelements then receiving lines 0 and 1 will be all that are employed.Similarly, for additional received diversity signals, additional outputsfrom function generator 36 will be used to supply control signalling tosignal selection switches 40 _(N) as dictated by the number of diversityreceiving elements contained in the receiver.

[0076] The signal received from an individual antenna is split between aphase adjusting or delay path 41 _(N) and a direct path 44 _(N) Delaypath 41 _(N) is provided to a delay line or phase changer 48 _(N) whichproduces an output signal on line 42 _(N) that is delayed by ½ cycle ofthe carrier frequency. Put another way, the phase of the carrier of thesignal on line 42 _(N) is shifted by pi radians. Signal selection switch40 _(N) is used to select between a delayed or phase shifted signal frompath 42 _(N) or a direct signal from path 44 _(N) for output on line 46_(N). The output selected from either path is selected in accordancewith the selection controlled by control line 38 _(N). The delay path 42_(N) provides the received signal which is delayed by 1/(2ƒ_(o)) secondsby a delay line implementation of phase changer 48 _(N), where θ_(o) isthe carrier frequency. Thus, the signal delay introduced by delay line48 _(N) corresponds to a carrier phase shift of n radians. There arevaried apparatus which may be advantageously employed to provide phasemodulating means 48 _(N) in accordance with the invention. A simpleinverting amplifier may be readily employed as a phase changer. Theinverting amplifier can have unity gain, or may be arranged to provideamplitude amplification of the received signal depending on therequirements of the application. Any changes to the amplitude of thesignal passing through For carrier frequencies that economically admitto it, a longer path may be switched with a shorter path to provide thenecessary phase delay. For example, at a carrier frequency in the 5gigahertz range, two paths which vary in length by a distance of 5 cmmay be used to provide a phase shift in accordance with the invention.

[0077] The circuit switching effected by signal selection switch 40 _(N)can be implemented using an arrangement of switching diodes or switchingtransistors as is well known to those skilled in the art. The outputselected by signal selection switch 40 _(N) is supplied on line 46 _(N)to summing circuit 50. Summing circuit 50 can be implemented as, forexample, an operational amplifier.

[0078] Function generator 36 will produce from 1 to N−1 outputs where Nrepresents the total number of receiving elements or antennas of thereceiver. In the preferred embodiment, where the receiver has 2antennas, function generator 36 is arranged to provide one output line38 a. It will be appreciated that an alternate configuration, which isnot the preferable configuration but is a working configuration, has thefirst antenna signal perturbed by W1 (shown in FIG. 3) and the second byW2 (of FIG. 3) or the first by W3 (of FIG. 3) and the second by W_(N) orall or any of these variations. While such a configuration operates inaccordance with the invention, the least cost and, therefore, preferableconfiguration is to avoid path switching one of the receivers.Naturally, If there are fewer than 2 antennas, then the spatialdiversity is not provided in the receiver and there is no need to have afunction generator 36.

[0079]FIG. 2a shows an alternate embodiment of the phase perturber andsignal combiner of FIG. 2. In this embodiment, all signal lines 1, 2, .. . N from the receiving elements are connected to an associatedswitched delay line 48 _(N) in contrast to the embodiment shown in FIG.2 which has one line 52 that directly connects to summer 50. Thealternate embodiment in this figure is depicted to illustrate that theinvention does not require one receiving element to be directlyconnected to the summer 50 to operate. From another aspect, it will beappreciated that this configuration of antennae 1, 2, . . . N andswitching elements 40 _(N) can be operated by function generator 30 insuch a fashion that one of the antenna, for example antenna 1, isconnected to summer 50 without ever switching in the associated phasechanger 48 a. In this manner of operation, the configuration of FIG. 2ais operationally identical to that of FIG. 2.

[0080] Referring to FIG. 3, timing diagrams for eight orthogonal outputsof function generator 36 in FIGS. 2 and 2a are shown. The functiongenerator outputs have symbol period transitions at the rate noted inthe column “Transitions per symbol period”. The symbol period for thereceiver is the span represented between the dotted lines T₀ and T₁during which time interval a single symbol will arrive at the receiver.During that symbol period, the control line 38 a corresponding-to thereceiving element 1 (of FIGS. 2 and 2a) will undergo the transitionclocking as shown for wave form W1. The control line for receiverelement 3 controlled by output on control line 38 b is shown as traceW2, and so on for each of the 8 traces shown in FIG. 2.

[0081] The timing diagrams depicted in FIG. 3 are derived from with thecharacteristics of the mutually orthogonal functions describedpreviously as expressed in equations P1, P2, P3 and P4 and the preferredfunctions are Walsh functions. The timing line traces of FIG. 3 may beexpanded to higher orders, if required, by referring to a more generalstatement of the requirements necessary of the functions which isprovided in the conditions outlined in formulas P1, P2, P3 and P4. Thetiming diagrams depicted in FIG. 3 are used to control the controlswitches 40 _(N) of FIGS. 2 and 2a. For each receiving element 0, 1, 2,3, . . . N, the carrier phase provided to summing circuit 50 isperturbed by associated delay line 48 _(N) switched in and out bycontrol switch 40 _(N), and the switching is controlled by control line38 _(N) which is operated by using a driving function output selectedfrom FIG. 3. The phase perturbation varies as a function of time and isgiven by: ${\psi_{i}(t)} = \{ \begin{matrix}0 & {;{{switch}\quad {on}\quad {right}\quad {contact}}} \\{\frac{1}{2f_{0}}{seconds}} & {;{{switch}\quad {on}\quad {left}\quad {contact}}}\end{matrix} $

[0082] where f₀ is the carrier frequency. In FIG. 3, the time between T₀and T₁ is selected to be a symbol time interval, or symbol period andthe control function outputs are mutually orthogonal over that timeperiod.

[0083] The traces in the timing diagram of FIG. 3 comply with theconditions defined by equations P1, P2, P3 and P4. Condition P1 appliesonly to a special case function which is a constant. Condition P3applies to a special case which is the comparison of a higher orderfunction with the constant function of condition P1 and, as such, is aspecific restatement of the more general condition expressed in P2.Condition P2 is satisfied by each trace of the timing diagram where eachtrace is high or one for one-half of the time and low or zero forone-half of the time. The requirement of conditions P2 and P3 that therebe an exact balance of output values between the high condition and lowcondition may be referred to as a fifty percent duty cycle requirement.The fifty percent duty cycle requirement is necessary to obtain mutualcancellation of the first and second cosine terms provided in the firstsummand term of E_(φ) given in equations (4) and (6). By inspection ofthe traces of FIG. 3, it will be seen that the condition defined byequation P4 is met between any two traces depicted in the timingdiagrams which may be referred to as the mutual orthogonalityrequirement. The mutual orthogonality requirement is necessary to obtainmutual cancellation of the four cosine terms provided in the secondsummand term of E_(φ) given in equations (4) and (6).

[0084] For clarity, the following table illustrates that waveform W7complies with condition P4 for each other waveform depicted namely: W1,W2, W3, W4, W5 and W6. To facilitate preparation of the table, thesymbol period is divided into 8 time slots numbered 1 through 8corresponding with the timing indicated by waveform W7. A high value ofeach waveform is depicted in the table as the value one and a low valueof each waveform is depicted in the table as the value zero. Thewaveform value pairs (a), (b), (c) and (d) correspond to the I and jfunction value pair labels provided in equation P4 and are shown in thecomparison row provided below each waveform pair being compared. TABLE 1Symbol Time Slot Waveform 1 2 3 4 5 6 7 8 W7 1 0 1 0 1 0 1 0 W6 1 0 1 00 1 0 1 W7 to W6 (d) (a) (d) (a) (b) (c) (b) (c) W7 1 0 1 0 1 0 1 0 W5 10 0 1 0 1 1 0 W7 to W5 (d) (a) (b) (c) (b) (c) (d) (a) W7 1 0 1 0 1 0 10 W4 1 0 0 1 1 0 0 1 W7 to W4 (d) (a) (b) (c) (d) (a) (b) (c) W7 1 0 1 01 0 1 0 W3 1 1 0 0 1 1 0 0 W7 to W3 (d) (c) (b) (a) (d) (c) (b) (a) W7 10 1 0 1 0 1 0 W2 1 1 0 0 0 0 1 1 W7 to W2 (d) (c) (b) (a) (b) (a) (d)(c) W7 1 0 1 0 1 0 1 0 W1 1 1 1 1 0 0 0 0 W7 to W1 (d) (c) (d) (c) (b)(a) (b) (a)

[0085] Thus as illustrated, waveform W7 meets condition P4 relative toall other waveforms W1 through W6 inclusive. As will be readilyunderstood, similar comparison tables may be made to compare waveforms:

[0086] W6 with W1, W2, W3, W4, W5 and W7

[0087] W5 with W1, W2, W3, W4, W6 and W7

[0088] W4 with W1, W2, W3, W5, W6 and W7

[0089] W3 with W1, W2, W4, W5, W6 and W7

[0090] W2 with W1, W3, W4, W5, W6 and W7

[0091] W1 with W2, W3, W4, W5, W6 and W7

[0092] to confirm that orthogonality condition P4 is met for each ofthese other waveforms relative to each other. Such further inspectionwill demonstrate that the waveforms depicted in FIG. 3 meet therequirements of condition P4. Consequently, the four cosine terms of thesecond summand given in equations (4) and (6) will mutually cancel eachother out.

[0093] It will be apparent that many changes and variations may be madeto the illustrative embodiments, while falling within the scope of theinvention and it is intended that all such changes and variations becovered by the claims appended hereto.

The embodiments of the invention in which an exclusive property orprivilege is claimed are defined as follows:
 1. Received signalprocessing apparatus for use with a spatial diversity receiver having afirst receiving element producing a received signal and at least onephysically separated diversity receiving element producing a receiveddiversity signal comprising: phase perturbation means for each receiveddiversity signal, to modulate the phase of the received diversity signalselectively by a predetermined phase adjustment in response to a controlsignal, producing a phase perturbed output signal; switching controlmeans to produce a control signal for each phase perturbation means; andsumming means to sum the received signal with all said phase perturbedoutput signals to produce a combined output signal.
 2. A receiver asclaimed in claim 1 wherein said switching control means produces controlsignals that are periodic over a symbol period.
 3. A receiver as claimedin claim 1 wherein said switching control means produces mutuallyorthogonal control signals.
 4. A receiver as claimed in claims 2 whereinsaid switching control means produces control signals selected from thefamily of functions called Walsh functions.
 5. A receiver as claimed inclaim 2 wherein said predetermined phase adjustment is substantiallyone-half the carrier cycle.
 6. A receiver as claimed in claim 2 whereinsaid phase perturbation means comprises a switched delay line path.
 7. Areceiver as claimed in claim 2 wherein said phase perturbation meanscomprises a switched negative unity gain amplifier.
 8. Received signalprocessing apparatus for use with a spatial diversity receiver having atleast two physically separated receiving elements each producing areceived signal comprising: phase perturbation means for each receivingelement to modulate the phase of the received signal selectively by apredetermined phase adjustment in response to a control signal,producing a phase perturbed output signal; switching control means toproduce a control signal for each phase perturbation means; and summingmeans to sum the outputs from all said phase perturbation means toproduce a combined signal output.
 9. A receiver as claimed in claim 8wherein said switching control means produces control outputs that areperiodic over a symbol period.
 10. A receiver as claimed in claim 9wherein said switching control means produces mutually orthogonalcontrol outputs.
 11. A receiver as claimed in claims 10 wherein saidswitching control means produces control outputs selected from thefamily of functions called Walsh functions.
 12. A receiver as claimed inclaims 8 wherein said predetermined phase adjustment is substantiallyone-half the carrier cycle.
 13. A receiver as claimed in claim 8 whereinsaid phase perturbation means comprises a switched delay line path. 14.A receiver as claimed in claim 8 wherein said phase perturbation meanscomprises a negative unity gain amplifier.
 15. A method for processingthe signals received by a diversity receiver having a first receivingelement and at least-one diversity receiving element comprising thesteps of: periodically perturbing the phase by a predetermined amount ofeach received signal produced by a diversity receiving element over asymbol period; and combining all received signals
 16. The method ofclaim 15 wherein the predetermined amount is substantially one-half ofthe carrier frequency.
 17. The method of claim 15 wherein said periodicperturbation has a duty cycle substantially equal to fifty percent ofthe symbol period.
 18. The method of claim 15 wherein each said periodicperturbation is mutually orthogonal to all others.
 19. A method forprocessing the signals received by a diversity receiver having at leasttwo receiving elements comprising the steps of: periodically perturbingeach received signal phase by a predetermined amount over a symbolperiod; and combining all received signals.
 20. The method of claim 15wherein the predetermined amount is substantially one-half of thecarrier frequency.
 21. The method of claim 19 wherein said periodicperturbation has a duty cycle substantially equal to fifty percent ofthe symbol period.
 22. The method of claim 20 wherein each said periodicperturbation is mutually orthogonal to all others.